Ensemble modem structure for imperfect transmission media

ABSTRACT

A high-speed modem that transmits and receives digital data on an ensemble of carrier frequencies spanning the usable band of a dial-up telephone line. The modem includes a system for variably allocating data and power among the carriers to compensate for equivalent noise and to maximize the data rate. Additionally, systems for eliminating the need for an equalization network, for adaptively allocating control of a channel, and the tracking variations in line parameters are disclosed.

This application is a division of Ser. No. 07/002,096 filed Jan. 12,1987, now U.S. Pat. No. 4,713,816 issued Mar. 15, 1988; which is adivision of Ser. No. 06/736,200 filed May 20, 1985, now U.S. Pat. No.4,679,227 issued July 7, 1987.

BACKGROUND OF THE INVENTION

1. Field of the Invention:

The invention relates generally to the field of data communications and,more particularly, to a high speed modem.

2. Description of the Prior Art:

Recently, specially designed telephone lines for the direct transmissionof digital data have been introduced. However, the vast majority oftelephone lines are designed to carry analog voice frequency (VF)signals. Modems are utilized to modulate VF carrier signals to encodedigital information on the VF carrier signals and to demodulate thesignals to decode the digital information carried by the signal.

Existing VF telephone lines have several limitations that degrade theperformance of modems and limit the rate at which data can betransmitted below desired error rates. These limitations include thepresence of frequency dependent noise on the VF telephone lines, afrequency dependent phase delay induced by the VF telephone lines, andfrequency dependent signal loss.

Generally, the usable band of a VF telephone line is from slightly abovezero to about four kHz. The power spectrum of the line noise is notuniformly distributed over frequency and is generally not determinative.Thus, there is no a priori method for determining the distribution ofthe noise spectrum over the usable bandwidth of the VF line.

Additionally, a frequency-dependent propagation delay is induced by theVF telephone line. Thus, for a complex multi-frequency signal, a phasedelay between the various components of the signal will be induced bythe VF telephone line. Again, this phase delay is not determinative andmust be measured for an individual VF telephone line at the specifictime that transmission takes place.

Further, the signal loss over the VF telephone line varies withfrequency. The equivalent noise is the noise spectrum component added tothe signal loss component for each carrier frequency, where bothcomponents are measured in decibels (dB).

Generally, prior art modems compensate for equivalent line noise andsignal loss by gear-shifting the data rate down to achieve asatisfactory error rate. For example, in U.S. Pat. 4,438,511, by Baran,a high speed modem designated SM9600 Super Modem manufactured by GandalfData, Inc., is described. In the presence of noise impairment, theSM9600 will "gear shift" or drop back its transmitted data rate to 4800bps or 2400 bps. The system described in the Baran patent transmits dataover 64 orthogonally modulated carriers. The Baran system compensatesfor the frequency dependent nature of the noise on the VF line byterminating transmission on carriers having the same frequency as thefrequency of large noise components on the line. Thus, Baran gracefullydegrades its throughput by ceasing to transmit on carrier frequencies atthe highest points of the VF line noise spectrum. The Baran systemessentially makes a go/no go decision for each carrier signal, dependingon the distribution of the VF line noise spectrum. This applicationreflects a continuation of the effort initiated by Baran.

Most prior art systems compensate for frequency dependent phase delayinduced by the VF line by an equalization system. The largest phasedelay is induced in frequency components near the edges of the usableband. Accordingly, the frequency components near the center of the bandare delayed to allow the frequency components at the outside of the bandto catch up. Equalization generally requires additional circuitry toaccomplish the above-described delays.

A further problem associated with two way transmission over the VFtelephone line is that interference between the outgoing and incomingsignals is possible. Generally, separation and isolation between the twosignals is achieved in one of three ways:

(a) Frequency multiplexing in which different frequencies are used forthe different signals. This method is common in modem-basedtelecommunication systems.

(b) Time multiplexing, in which different time segments are used for thedifferent signals. This method is often used in half-duplex systems inwhich a transmitter relinquishes a channel only after sending all thedata it has. And,

(c) Code multiplexing, in which the signals are sent using orthogonalcodes.

All of the above-described systems divide the space available accordingto constant proportions fixed during the initial system design. Theseconstant proportions, however, may not be suitable to actual trafficload problem presented to each modem. For example, a clerk at a PC workstation connected to a remote host computer may type ten or twentycharacters and receive a full screen in return. In this case, constantproportions allocating the channel equally between the send and receivemodems would greatly overallocate the channel to the PC work stationclerk. Accordingly, a modem that allocates channel capacity according tothe needs of the actual traffic load situation would greatly increasethe efficient utilization of the channel capacity.

SUMMARY OF THE INVENTION

The present invention is a high-speed modem for use with dial-up VFtelephone lines. The modem utilizes a multicarrier modulation scheme andvariably allocates data and power to the various carriers to maximizethe overall data transmission rate. The allocation of power among thecarriers is subject to the constraint that the total power allocatedmust not exceed a specified limit.

In a preferred embodiment, the modem further includes a variableallocation system for sharing control of a communication link betweentwo modems (A and B) according to actual user requirements.

Another aspect of the invention is a system for compensating forfrequency dependent phase delay and preventing intersymbol interferencethat does not require an equalization network.

According to one aspect of the invention, quadrature amplitudemodulation (QAM) is utilized to encode data elements of varyingcomplexity on each carrier. The equivalent noise component at eachcarrier frequency is measured over a communication link between twomodems (A and B).

As is known in the art, if the bit error rate (BER) is to be maintainedbelow a specified level, then the power required to transmit a dataelement of a given complexity on a given carrier frequency must beincreased if the equivalent noise component at that frequency increases.Equivalently, to increase data complexity, the signal to noise ratio,S/N, must be increased.

In one embodiment of the present invention, data and power are allocatedto maximize the overall data rate within external BER and totalavailable power constraints. The power allocation system computes themarginal required power to increase the symbol rate on each carrier fromn to n+1 information units. The system then allocates information unitsto the carrier that requires the least additional power to increase itssymbol rate by one information unit. Because the marginal powers aredependent on the values of the equivalent noise spectrum of theparticular established transmission link, the allocation of power anddata is specifically tailored to compensate for noise over thisparticular link.

According to another aspect of the invention, a first section of thesymbol on each carrier is retransmitted to form a guard-time waveform ofduration T_(E) +T_(PH) where T_(E) is the duration of the symbol andT_(PH) is the duration of the first section. The magnitude of T_(PH) isgreater than or equal to the maximum estimated phase delay for anyfrequency component of the waveform. For example, if the symbol isrepresented by the time series, x₀ . . . x_(n-1), transmitted in timeT_(E) ; then the guardtime waveform is represented by the time series,x₀ . . . x_(n-1), x₀ . . . x_(m-1), transmitted in time T_(E) +T_(PH).The ratio that m bears to n is equal to the ratio that T_(PH) bears toT_(E).

At the receiving modem, the time of arrival, T₀, of the first frequencycomponent of the guard-time waveform is determined. A sampling period,of duration T_(E), is initiated a time T₀ +T_(PH).

Accordingly, the entire symbol on each carrier frequency is sampled andintersymbol interference is eliminated.

According to a still further aspect of the invention, allocation ofcontrol to the transmission link between modems A and B is accomplishedby setting limits to the number of packets that each modem may transmitduring one transmission cycle. A packet of information comprises thedata encoded on the ensemble of carriers comprising one waveform. Eachmodem is also constrained to transmit a minimum number of packets tomaintain the communication link between the modems. Thus, even if onemodem has no data to transmit, the minimum packets maintain timing andother parameters are transmitted. On the other hand, if the volume ofdata for a modem is large, it is constrained to transmit only themaximum limited number of packets, N, before relinquishing control tothe other modem.

In practice, if modem A has a small volume of data and modem B has alarge volume of data, modem B will have control of the transmission linkmost of the time. If control is first allocated to modem A it will onlytransmit the minimal number, I, of packets. Thus A has control for onlya short time. Control is then allocated to B which transmits N packets,where N may be very large. Control is again allocated to modem A whichtransmits I packets before returning control to B.

Thus, allocation of control is proportional to the ratio of I to N. Ifthe transmission of the volume of data on modem A requires L packets,where L is between I and N, then the allocation is proportional to theratio of L to N. Accordingly, allocation of the transmission link variesaccording to the actual needs of the user.

Additionally, the maximum number of packets, N, need not be the same foreach modem, but may be varied to accommodate known disproportions in thedata to be transmitted by A and B modems.

According to another aspect of the invention, signal loss and frequencyoffset are measured prior to data determination. A tracking systemdetermines variations from the measured values and compensates for thesedeviations.

According to a further aspect of the invention, a system for determininga precise value of T₀ is included. This system utilizes two timingsignals, at f₁ and f₂, incorporated in a waveform transmitted from modemA at time T_(A). The relative phase difference between the first andsecond timing signals at time T_(A) is zero.

The waveform is received at modem B and a rough estimate, T_(EST), ofthe time of reception is obtained by detecting energy at f₁. Therelative phase difference between the timing signals at time T_(EST) isutilized to obtain a precise timing reference, T₀.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a graph of the ensemble of carrier frequencies utilized in thepresent invention.

FIG. 2 is a graph of the constellation illustrating the QAM of eachcarrier.

FIG. 3 is a block diagram of an embodiment of the invention.

FIG. 4 is a flow chart illustrating the synchronization process of thepresent invention.

FIG. 5 is a series of graphs depicting the constellations for 0, 2, 4,5, 6 bit data elements and exemplary signal to noise ratios and powerlevels for each constellation.

FIG. 6 is a graph illustrating the waterfilling algorithm.

FIG. 7 is a histogram illustrating the application of the waterfillingalgorithm utilized in the present invention.

FIG. 8 is a graph depicting the effects of phase dependent frequencydelay on frequency components in the ensemble.

FIG. 9 is a graph depicting the wave forms utilized in the presentinvention to prevent intersymbol interference.

FIG. 10 is a graph depicting the method of receiving the transmittedensemble.

FIG. 11 is a schematic diagram depicting the modulation template.

FIG. 12 is a schematic diagram depicting the quadrants of one square inthe modulation template.

FIG. 13 is a schematic diagram of a hardware embodiment of the presentinvention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

The present invention is a modem that adaptively allocates power betweenvarious carrier frequencies in a frequency ensemble to compensate forfrequency dependent line noise, eliminates the need for equalizationcircuitry to compensate for a frequency dependent phase delay, andprovides a duplex mechanism that accounts for varying channel loadconditions to allocate the channel between the send and receive modems.Additional features of the invention are described below.

A brief description of the frequency ensemble and modulation schemeutilized in the present invention is first presented with respect toFIGS. 1 and 2 to facilitate the understanding of the invention. Aspecific embodiment of the invention is then described with reference toFIG. 3. Finally, the operation of various features of the invention aredescribed with reference to FIGS. 4 through 13.

Modulation and Ensemble Configuration

Referring now to FIG. 1, a diagrammatic representation is shown of thetransmit ensemble 10 of the present invention. The ensemble includes 512carrier frequencies 12 equally spaced across the available 4 kHz VFband. The present invention utilizes quadrature amplitude modulation(QAM) wherein phase independent sine and cosine signals at each carrierfrequency are transmitted. The digital information transmitted at agiven carrier frequency is encoded by amplitude modulating theindependent sine and cosine signals at that frequency.

The QAM system transmits data at an overall bit rate, R_(B). However,the transmission rate on each carrier, denoted the symbol or baud rate,R_(S), is only a fraction of R_(B). For example, if data were allocatedequally between two carriers then R_(S) =R_(B) /2.

In the preferred embodiment 0, 2, 4, 5 or 6 bit data elements areencoded on each carrier and the modulation of each carrier is changedevery 136 msec. A theoretical maximum, R_(B), assuming a 6 bit R_(S) foreach carrier, of 22,580 bit/sec (bps) results. A typical relizableR_(S), assuming 4 bit R_(S) over 75% of the carriers, is equal to about11,300 bps. This extremely high R_(S) is achieved with a bit error rateof less than 1 error/100,000 bits transmitted.

In FIG. 1, a plurality of vertical lines 14 separates each ensemble intotime increments known hereafter as "epochs." The epoch is of durationT_(E) where the magnitude of T_(E) is determined as set forth below.

The QAM system for encoding digital data onto the various carrierfrequencies will now be described with reference to FIG. 2. In FIG. 2 afour bit "constellation" 20 for the nth carrier is depicted. A four bitnumber may assume sixteen discrete values. Each point in theconstellation represents a vector (x_(n), y_(n)) with x_(n) being theamplitude of the sine signal and y_(n) being the amplitude of the cosinesignal in the above-described QAM system. The subscript n indicates thecarrier being modulated. Accordingly, the four bit constellationrequires four discrete y_(n) and four discrete x_(n) values. Asdescribed more fully below, increased power is required to increase thenumber of bits transmitted at a given carrier frequency due to theequivalent noise component at that frequency. The receive modem, in thecase of four bit transmission, must be able to discriminate between fourpossible values of the x_(n) and y_(n) amplitude coefficients. Thisability to discriminate is dependent on the signal to noise ratio for agiven carrier frequency.

In a preferred embodiment, packet technology is utilized to reduce theerror rate. A packet includes the modulated epoch of carriers and errordetection data. Each packet in error is retransmitted until correct.Alternatively, in systems where retransmission of data is undesirable,epochs with forward error correcting codes may be utilized.

Block Diagram

FIG. 3 is a block diagram of an embodiment of the present invention. Thedescription that follows is of an originate modem 26 coupled to anoriginate end of a communication link formed over a public switchedtelephone line. It is understood that a communication system alsoincludes an answer modem coupled to the answer end of the communicationlink. In the following discussion, parts in the answer modemcorresponding to identical or similar parts in the originate modem willbe designated by the reference number of the originate modem primed.

Referring now to FIG. 3, an incoming data stream is received by a sendsystem 28 of the modem 26 at data input 30. The data is stored as asequence of data bits in a buffer memory 32. The output of buffer memory32 is coupled to the input of a modulation parameter generator 34. Theoutput of the modulation parameter generator 34 is coupled to a vectortable buffer memory 36 with the vector table buffer memory 36 alsocoupled to the input of a modulator 40. The output of the modulator 40is coupled to a time sequence buffer 42 with the time sequence buffer 42also coupled to the input of a digital-to-analog converter 43 includedin an analog I/O interface 44. The interface 44 couples the output ofthe modem to the public switched telephone lines 48.

A receive system 50 includes an analog-to-digital converter (ADC) 52coupled to the public switched telephone line 48 and included in theinterface 44. The output from the ADC 52 is coupled to a receive timeseries buffer 54 which is also coupled to the input of a demodulator 56.The output of the demodulator 56 is coupled to a receive vector tablebuffer 58 which is also coupled to the input of a digital data generator60. The digital data generator 60 has an output coupled to a receivedata bit buffer 62 which is also coupled to an output terminal 64.

A control and scheduling unit 66 is coupled with the modulationparameter generator 34, the vector table buffer 36, the demodulator 56,and the receive vector table buffer 58.

An overview of the functioning of the embodiment depicted in FIG. 3 willnow be presented. Prior to the transmission of data, the originate modem26, in cooperation with the answer modem 26', measures the equivalentnoise level at each carrier frequency, determines the number of bits perepoch to be transmitted on each carrier frequency, and allocates powerto each carrier frequency as described more fully below.

The incoming data is received at input port 30 and formatted into a bitsequence stored in the input buffer 32.

The modulator 34 encodes a given number of bits into an (x_(n), y_(n))vector for each carrier frequency utilizing the QAM system describedabove. For example, if it were determined that four bits were to betransmitted at frequency f_(n) then four bits from the bit stream wouldbe converted to one of the sixteen points in the four bit constellationof FIG. 2. Each of these constellation points corresponds to one ofsixteen possible combinations of four bits. The amplitudes of the sineand cosine signals for frequency n then corresponds to the point in theconstellation encoding the four bits of the bit sequence. The (x_(n),y_(n)) vectors are then stored in the vector buffer table 36. Themodulator receives the table of (x_(n), y_(n)) vectors for the carriersin the ensemble and generates a digitally encoded time seriesrepresenting a wave form comprising the ensemble of QAM carrierfrequencies.

In a preferred embodiment the modulator 40 includes a fast Fouriertransform (FFT) and performs an inverse FFT operation utilizing the(x,y) vectors as the FFT coefficients. The vector table includes 1,024independent points representing the 1,024 FFT points of the 512frequency constellation. The inverse FFT operation generates 1,024points in a time series representing the QAM ensemble. The 1,024elements of this digitally encoded time series are stored in the digitaltime series buffer 42. The digital time sequence is converted to ananalog wave form by the analog to digital converter 43 and the interface46 conditions the signal for transmission over the public switchedtelephone lines 48.

Turning now to the receive system 50, the received analog waveform fromthe public switched telephone lines 48 is conditioned by the interface46 and directed to the analog to digital converter 52. The analog todigital converter 52 converts the analog waveform to a digital 1,024entry time series table which is stored in the receive time seriesbuffer 54. The demodulator 56 converts the 1,024 entry time series tableinto a 512 entry (x_(n), y_(n)) vector table stored in the receivevector table buffer 58. This conversion is accomplished by performing anFFT on the time series. Note that information regarding the number ofbits encoded onto each frequency carrier has been previously stored inthe demodulator and digital data generator 60 so that the (x,y) tablestored in the receive vector table buffer 58 may be transformed to anoutput data bit sequence by the digital data generator 60. For example,if the (x_(n), y_(n)) vector represents a four bit sequence then thisvector would be converted to a four bit sequence and stored in thereceive data bit buffer 62 by the digital data generator 60. The receivedata bit sequence is then directed to the output 64 as an output datastream.

A full description of the FFT techniques utilized is described in a bookby Rabiner et al., entitled Theory and Applications of Digital SignalProcessing, Prentice-Hall, Inc., N.J., 1975. However, the FFT modulationtechnique described above is not an integral part of the presentinvention. Alternatively, modulation could be accomplished by directmultiplication of the carrier tones as described in the above-referencedBaran patent, which is hereby incorporated by reference, at col. 10,lines 13-70, and col. 11, lines 1-30. Additionally, the demodulationsystem described in Baran at col. 12, lines 35-70, col. 13, lines 1-70,and col. 14, lines 1-13 could be substituted.

The control and scheduling unit 66 maintains overall supervision of thesequence of operations and controls input and output functions.

Determination of Equivalent Noise

As described above, the information content of the data element encodedon each frequency carrier and the power allocated to that frequencycarrier depends on the magnitude of the channel noise component at thatcarrier frequency. The equivalent transmitted noise component atfrequency f_(n), N(f_(n)), is the measured (received) noise power atfrequency f_(n) multiplied by the measured signal loss at frequencyf_(n). The equivalent noise varies from line to line and also varies ona given line at different times. Accordingly, in the present system,N(f) is measured immediately prior to data transmission.

The steps of a synchronization technique utilized in the present systemto measure N(f) and establish a transmission link between answer andoriginate modems 26 and 26' are illustrated in FIG. 4. Referring now toFIG. 4, in step 1 the originate modem dials the number of the answermodem and the answer modem goes off hook. In step 2 the answer modemtransmits an epoch of two frequencies at the following power levels:

(a) 1437.5 Hz. at -3 dBR; and

(b) 1687.5 Hz at -3 dBR.

The power is measured relative to a reference, R, where, in a preferredembodiment, OdBR=-9 dBm, m being a millivolt. These tones are used todetermine timing and frequency offset as detailed subsequently.

The answer modem then transmits an answer comb containing all 512frequencies at -27 dBR. The originate modem receives the answer comb andperforms an FFT on the comb. Since the power levels of the 512frequencies were set at specified values, the control and schedulingunit 66 answer modem 26 compares the (x_(n),y_(n)) values for eachfrequency of the received code and compares those values to a table of(x_(n),y_(n)) values representing the power levels of the transmittedanswer code. This comparison yields the signal loss at each frequencydue to the transmission over the VF telephone lines.

During step 3 both the originate and answer modems 26 and 26' accumulatenoise data present on the line in the absence of any transmission byeither modem. Both modems then perform an FFT on the accumulated noisesignals to determine the measured (received) noise spectrum componentvalues at each carrier frequency. Several epochs of noise may beaveraged to refine the measurement.

In step 4 the originate modem transmits an epoch of two frequenciesfollowed by an originate comb of 512 frequencies with the same powerlevels described above for step 2. The answer modem receives the epochand the originate comb and calculates the timing, frequency offset andsignal loss values at each carrier frequency as described above for theoriginate modem in step 2. At this point the originate modem 26 hasaccumulated noise and signal loss data for transmission in the answeroriginate direction while the answer modem has accumulated the same datarelating to transmission in the originate answer direction. Each modemrequires data relating to transmission loss and receive noise in boththe originate-answer and answer-originate directions. Therefore, thisdata is exchanged between the two modems according to the remainingsteps of the synchronization process.

In step 5 the originate modem generates and transmits a first phaseencoded signal indicating which carrier frequencies will support two bittransmission at standard power levels in the answer-originate direction.Each component that will support two bits in the answer-originatedirection at a standard power level is generated as a -28 dBR signalwith 180° relative phase. Each component that will not support two bittransmission in the answer-originate direction at the standard powerlevel is coded as a -28 dBR, 0° relative phase signal. The answer modemreceives this signal and determines which frequency carriers willsupport two bit transmission in the answer-originate direction.

In step 6 the answer modem generates and transmits a second phaseencoded signal indicating which carrier frequencies will support two bittransmission in both the originate-answer and answer-originatedirections. The generation of this signal is possible because the answermodem has accumulated noise and signal loss data in the originate-answerdirection and has recieved the same data for the answer-originatedirection in the signal generated by the originate modem in step 5. Inthe signal generated by the originate modem, each frequency componentthat will support two bits in both directions is coded with 180°relative phase and all other components are coded with 0° relativephase.

A transmission link now exists between the two modems. In general, 300to 400 frequency components will support two bit transmission at astandard power level, thereby establishing about a 600 bit/epoch ratebetween the two modems. In step 7 the originate modem sends data on thenumber of bits (0 to 15) and the power levels (0 to 63 dB) that can besupported on each frequency in the answer-originate direction inensemble packets formed over this existing data link. Accordingly, boththe originate and answer modem now have the data relating totransmission in the answer-originate direction. The steps forcalculating the number of bits and power levels that can be supported oneach frequency component will be described below.

In step 8 the answer modem sends data on the number of bits and powerlevels that can be supported on each frequency in the originate-answerdirection utilizing the existing data link. Thus, both modems areapprised of the number of bits and power levels to be supported on eachfrequency component in both the answer-originate and originate-answerdirections.

The above description of the determination of the equivalent noise levelcomponent at each carrier frequency sets forth the required steps in agiven sequence. However, the sequence of steps is not critical and manyof the steps may be done simultaneously or in different order, forexample, the performance of the FFT on the originate code and theaccumulation of noise data may be done simultaneously. A precise timingreference is also calculated during the synchronization process. Thecalculation of this timing reference will be described more fully belowafter the description of the method for calculating the number of bitsand power levels allocated to each frequency component.

It is a common VF telephone line impairment that a frequency offset, ofup to 7 Hz, exists between transmitted and received signals. This offsetmust be corrected for the FFT to function reliably. In a preferredembodiment, this correction is achieved by performing a single sidebandmodulation of the quadrature tones at the offset frequency by the trueand Hilbert images of received signal. Synchronization and trackingalgorithms generate estimates of the frequency offset necessary.

Power and Code Complexity Allocation

The information encoded on each carrier frequency signal is decoded atthe receiver channel by the demodulator 56. Channel noise distorts thetransmitted signal and degrades the accuracy of the demodulationprocess. The transmission of a data element having a specifiedcomplexity, e.g., B₀ bits at a specified frequency, f₀, over a VFtelephone line characterized by an equivalent noise level component, N₀,will now be analyzed. Generally, external system requirements determinea maximum bit error rate (BER) that can be tolerated. For thetransmission of b₀ bits at noise level N₀ and frequency f₀, the signalto noise ratio must exceed E_(b) /N₀ where E_(b) is the signal power perbit to maintain the BER below a given BER, (BER)₀.

FIG. 5 depicts the QAM constellations for signals of variouscomplexities B. An exemplary signal to noise ratio, E_(b) /N₀, for eachconstellation and the power required to transmit the number of bits inthe constellation without exceeding (BER)₀ is depicted alongside eachconstellation graph.

A modem operates under the constraint that the total available powerplaced on the public switched telephone lines may not exceed a value,P₀, set by the telephone companies and government agencies. Thus, signalpower may not be increased indefinitely to compensate for line noise.Accordingly, as noise increases, the complexity of the signalstransmitted must be decreased to maintain the required BER.

Most existing modems arbitrarily gear shift the signal complexity downas line noise power increases. For example, one prior art modem reducesthe transmitted data rate from a maximum of 9,600 bps to steps of 7,200bps, 4,800 bps, 2,400 bps, 1,200 bps, and so on until the bit error rateis reduced below a specified maximum. Accordingly, the signal rate isdecreased in large steps to compensate for noise. In the Baran patent,the method for reducing the transmission rate takes into account thefrequency dependent nature of the noise spectrum. There, each channelcarries a preset number of bits at a specified power level. The noisecomponent at each frequency is measured and a decision is made whetherto transmit at each carrier frequency. Thus, in Baran, the data ratereduction scheme compensates for the actual distribution of the noiseover the available bandwidth.

In the present invention, the complexity of the signal on each frequencycarrier and the amount of the available power allocated to eachfrequency carrier is varied in response to the frequency dependence ofthe line noise spectrum.

The present system for assigning various code complexities and powerlevels to the frequency component signals in the ensemble is based onthe waterfilling algorithm. The waterfilling algorithm is an informationtheroretic way of assigning power to a channel to maximize the flow ofinformation across the channel. The channel is of the type characterizedby an uneven noise distribution and the transmitter is subject to apower constraint. FIG. 6 provides a visualization of the waterfillingalgorithm. Referring now to FIG. 6, power is measured along the verticalaxis and frequency is measured along the horizontal axis. The equivalentnoise spectrum is represented by the solid line 70 and the availablepower is represented by the area of the cross hatched region 72. Thename waterfilling comes from the analogy of the equivalent noisefunction to a series of valleys in a mountain filled with a volume ofwater representing the assigned power. The water fills the valleys andassumes a level surface. A theoretical description of the waterfillingalgorithm is given in the book by Gallagher, entitled Information TheoryAnd Reliable Communication; J. Wiley and Sons, New York, 1968, p. 387.

It must be emphasized that the waterfilling theorem relates tomaximizing the theoretical capacity of a channel where the capacity isdefined as the maximum of all data rates achievable using differentcodes, all of which are error correcting, and where the best tend to beof infinite length.

The method utilizing the present invention does not maximize thecapacity of the channel. Instead, the method maximizes the amount ofinformation transmitted utilizing the QAM ensemble described above withrespect to FIG. 1 and subject to an available power restriction.

An implementation of the waterfilling concept is to allocate anincrement of available power to the carrier having the lowest equivalentnoise floor until the allocated power level reaches the equivalent noiselevel of the second lowest carrier. This allocation requires a scanthrough the 512 frequencies.

Incremental power is then allocated between the lowest two carriersuntil the equivalent noise level of the third lowest channel is reached.This allocation level requires many scans through the frequency tableand is computationally complex.

The power allocation method used in a preferred embodiment of thepresent invention is as follows:

(1) Calculate the system noise at the transmitter by measuring theequivalent noise at the receiver and multiplying by transmission loss.This process for measuring these quantities was described above withrespect to synchronization and FIG. 4. The system noise components arecalculated for each carrier frequency.

(2) For each carrier frequency, calculate the power levels required totransmit data elements of varying complexity (in the present case, 0, 2,4, 5, 6, and 8 bits). This is accomplished by multiplying the equivalentnoise by the signal to noise ratios necessary for transmission of thevarious data elements with a required BER, for example one error per100,000 bits. The overall BER is the sum of the signal error rates ofeach modulated carrier. These signal to noise ratios are available fromstandard references, and are wellknown in the art.

(3) From the calculated required transmission power levels, the marginalrequired power levels to increase data element complexity aredetermined. These marginal required power levels are the difference intransmission power divided by the quantitative difference in complexityof the data elements closest in complexity.

(4) For each channel generate a two column table of marginal requiredpower levels and quantitative differences where the units are typicallyexpressed as Watts and bits, respectively.

(5) Construct a histogram by organizing the table of step 4 according toincreasing marginal power.

(6) Assign the available transmitter power sequentially over theincreasing marginal powers until available power is exhausted.

The power allocation method may be better understood through a simpleexample. The numbers presented in the example are not intended torepresent parameters encountered in an operating system.

Table 1 sets out the power requirement, P, to transmit a data element ofa selected number of bits, N₁, for two carriers A and B at frequenciesf_(A) and f_(B).

                  TABLE 1                                                         ______________________________________                                        N.sub.1                                                                              N.sub.2 -N.sub.1                                                                           P     MP(N.sub.1 to N.sub.2)                              ______________________________________                                        Carrier A                                                                     0      --            0    --                                                  2      2             4    MP(0 to 2)=2/bit                                    4      2            12    MP(2 to 4)=4/bit                                    5      1            19    MP(4 to 5)=7/bit                                    6      1            29     MP(5 to 6)=10/bit                                  Carrier B                                                                     0      --            0    --                                                  2      2             6    MP(0 to 2)=3/bit                                    4      2            18    MP(2 to 4)=6/bit                                    5      1            29     MP(4 to 5)=11/bit                                  6      1            44     MP(5 to 6)=15/bit                                  ______________________________________                                    

The marginal power to increase the complexity from a first number ofbits, N₁, to a second number of bits, N₂, is defined by therelationship: ##EQU1## where P₂ and P₁ are the powers required totransmit data elements of complexity N₂ and N₁. N₂ -N₁ is quantitativedifference in the complexity of the data elements. It is understood theBER is constrained to remain below a preset limit.

The marginal powers for f_(A) are less than for f_(B) because theequivalent noise at f_(B), N(f_(B)), is greater than the equivalentnoise at f_(A), N(f_(A)).

The implementation of the allocation scheme for carriers A and B willnow be described. Assume that a total number of bits, N_(T), are encodedon the ensemble but that no bits have been assigned to carriers A or B.For example, N(f_(A)) and N(f_(A)) might be greater than the powers ofthose carriers already carrying the data.

In this example, the system is to allocate ten remaining available powerunits between carriers A and B to increase the overall data elementcomplexity by the maximum amount.

To increase N_(T) by two bits requires that four units of power beallocated if channel A is utilized and that six units of power beallocated in channel B is utilized. This follows because for bothchannels N₁ =0 and N₂ =2 and MP(0 to 2)=2/bit for channel A and MP(0 to2)=3/bit for channel B. Therefore, the system allocates four units ofpower to carrier A, encodes a two bit data element on carrier A,increases the overall signal complexity from N_(T) to N_(T) +2, and hassix remaining available power units.

The next increase of two bits requires six power units because MP(2 to4)=4/bit for carrier A and MP(0 to 2)=3/bit for channel B. Therefore,the system allocates six units of power to carrier B, encodes a two bitdata element on carrier B, increases the overall signal complexity fromN_(T) +2 to N_(T) +4 bits, and has no remaining available power units.

As is now clear, the system "shops" among the various carrierfrequencies for the lowest power cost to increase the complexity of theoverall ensemble data element.

The allocation system is extended to the full 512 carrier ensemble byfirst generating the tables of Table 1 for each carrier during a firstpass through the frequencies.

A histogram organizing the calculated marginal required power levels forall the carriers according to increasing power is then constructed. FIG.7 is a depiction of an exemplary histogram constructed according to thepresent method.

In FIG. 7 the entire table of marginal powers is not displayed. Instead,the histogram is constructed having a range of 64 dB with counts spacedin 0.5 dB steps. The quantitative differences between the steps areutilized as counts. Although this approach results in a slight round-offerror, a significant reduction in task length is achieved. The methodused to construct the histogram is not critical to practicing theinvention.

Each count of the histogram has an integer entry representing the numberof carriers having a marginal power value equal to the power value atthe count. The histogram is scanned from the lowest power level. Theinteger entry at each count is multiplied by the number of counts andsubtracted from the available power. The scan continues until availablepower is exhausted.

When the scan is completed it has been determined that all marginalpower values below a given level, MP(max), are acceptable for power anddata allocation. Additionally, if available power is exhausted partiallythrough marginal power level, MP(max), then k additional carriers willbe allocated power equal to MP(max+1).

The system then scans through the ensemble again to allocate power anddata to the various carriers. The amount of power allocated to eachcarrier is the sum of marginal power values for that carrier less thanor equal to MP(max). Additionally, an amount of power equal to MP(max+1)will be allocated if the k MP(max+1) values have not been previouslyallocated.

Timing and Phase Delay Compensation

The reconstruction of (x,y) vector table by the receive system requires1024 time samples of the received waveform. The bandwidth is about 4 kHzso that Nyquist sampling rate about 8000/sec and the time sample offsetbetween samples is 125 microseconds. The total sampling time is thus 128msec. Similarly, the transmit FFT generates a time series having 1024entries and the symbol time is 128 msec.

The sampling process requires a timing reference to initiate thesampling. This timing reference is established during synchronization bythe following method:

During the synchronization steps defined with reference to FIG. 4, theoriginate modem detects energy at the 1437.5 Hz frequency component (thefirst timing signal) in the answer comb at time T_(EST). This time is arough measure of the precise time that the first timing frequencycomponent arrives at the receiver and is generally accurate to about 2msec.

This rough measure is refined by the following steps. The first timingsignal and a second timing signal (at 1687.5 Hz) are transmitted withzero relative phase at the epoch mark.

The originate modem compares the phases of the first and second timingsignals at time T_(EST). The 250 Hz frequency difference between thefirst and second timing signals results in an 11° phase shift betweenthe two signals for each 125 microsecond time sample offset. The firstand second timing signals have low relative phase distortion (less than250 microseconds) due to their location near the center of the band.Accordingly, by comparing the phases of the two timing samples andcorrecting T_(EST) by the number of time sampling offsets indicated bythe phase difference, a precise timing reference, T₀, can be determined.

A further difficulty relating to timing the sampling process relates tofrequency dependent phase delay induced by the VF line. This phase delaytypically is on the order of 2 msec, or more, for VF telephone lines.Further, this phase delay is significantly worse near the edges of the 4kHz usable band.

FIG. 8 depicts distribution of the frequency carriers of the ensembleafter undergoing frequency dependent phase delay. Referring to FIG. 8,three signals 90, 92, and 94 at frequencies f₀, f₂₅₆, and f₅₁₂ aredepicted. Two symbols, x_(i) and y_(i), of length T_(S) are transmittedat each frequency. Note that the duration of each symbol is not changed.However, the leading edge of the signals near the edge of the band 92and 94 are delayed relative to those signals near the center of the band94.

Additionally, for two sequentially transmitted epochs x_(i) and y_(i)the trailing section of the first symbol x_(i) on signals 92 and 96,near the outer edge of the band will overlap the leading edge of thesecond symbol y_(i) on the signal 94 near the center of the band. Thisoverlap results in intersymbol interference.

If the sampling interval is framed to sample a given time interval,T_(S), then complete samples of every carrier in the ensemble will notbe obtained and signals from other epochs will also be sampled.

Existing systems utilize phase correction (equalization) networks tocorrect for phase distortion and to prevent intersymbol interference.

The present invention utilizes a unique guard-time format to eliminatethe need for an equalization network. This format is illustrated in FIG.9.

Referring now to FIG. 9, first, second, and third transmitted symbols,represented by time series x_(i), y_(i), and z_(i), respectively, aredepicted. The waveforms depicted in FIG. 3 are modulated on one of thecarriers at frequency f. In this example a symbol time, T_(S), of 128msec. and a maximum phase delay, T_(PH), of 8 msec are assumed. Aguard-time waveform is formed by repeating the first 8 msec. of thesymbol. The guard-time waveform defines an epoch of 136 msec. Forexample, in the first waveform 110, (X_(i)), the time series of thesymbol, X₀ -X₁₀₂₃, is first transmitted, then the first 8 msec. of thesymbol, X₀ -X₆₃, are repeated.

The sampling of the epoch is aligned with the last 128 msec. of theguard-time waveform (relative to the beginning of the guard-time epochdefined by those frequency components which arrive first).

This detection process is illustrated in FIG. 10. In FIG. 10 first andsecond guard-time waveforms 110 and 112 at f₁, near the center of theband, and f₂, near the edge of the band, are depicted. The frequencycomponent at f₁ is the component of the ensemble that arrives first atthe receiver and the component at f₂ arrives last. In FIG. 10 the secondwaveform 112, at f₂, arrives at the receiver at T₀ +T_(PH), which is 8msec. after the time, T₀, that the first waveform 110, at f₁, arrives atthe receiver. The sampling period of 128 msec. is initiated at the timeT₀ +T_(PH). Thus, the entire symbol on f₂, X₀ -X₁₀₂₃, is sampled. Theentire symbol at f₁ is also sampled because the initial 8 msec. of thatsymbol has been retransmitted.

Also, intersymbol interference has been eliminated. The arrival of thesecond symbol, (y_(i)), at f₁ has been delayed 8 msec. by theretransmission of the first 8 msec. of (x_(i)). Thus, the leading edgeof the second symbol at f₁, does not overlap the trailing edge of thefirst symbol at f₂.

The 8 msec. guardtime reduces the usable time-bandwidth product of thesystem by only about 6%. This small decrease is due to the very longduration of each symbol relative to the necessary guardtime.

Tracking

In pratice, for a given carrier, the magnitudes of the (x,y) vectorsextracted during the demodulation process do not fall exactly at theconstellation points but are distributed over a range about each pointdue to noise and other factors. Accordingly, the signal is decodedutilizing a modulation template as depicted in FIG. 11.

Referring now to FIG. 11, the template is formed by a grid of squares113 with the constellation points 114 at the centers of the squares 113.

In FIG. 11, the vector W=(x_(n),y_(n)) represents the demodulatedamplitudes of the sine and cosine signals at f_(n). W is in the square113 having the constellation point (3,3) centered therein. Accordingly,W is decoded as (3,3).

The present invention includes a system for tracking to determinechanges in transmission loss, frequency offset, and timing from thevalues determined during synchronization.

This tracking system utilizes the position of the received vectors inthe squares of the demodulation template of FIG. 11. In FIG. 12, asingle square is divided into four quadrants upper left, lower right,upper right, lower right, 115, 116, 117, and 118 characterized as toofast, too slow, too big, and too little, respectively. If counts in allfour quadrants over time by frequency or over frequency at one time areequal or nearly equal then the system is in alignment. That is, if noiseis the only impairment, then the direction of error for the decodedvector, W, should be random.

However, if transmission loss changes by even 0.1 dB the number of toosmall counts will vary significantly from the number of too largecounts. Similarly, a large difference between the number of too fast andtoo slow counts indicates a phase rotation caused by a change in theoffset frequency. Thus, the differences between the too fast, too slow,and too big, too small counts is an error characteristic that tracksvariations in signal loss and offset frequency.

The present invention utilizes this error characteristic to adjust thesignal loss and frequency offset determined during synchronization. Foreach frequency an adjustment of ±0.1 dB or ±1.0° is made depending onthe error characteristic. Other divisions of the decoding region intodistinct or overlapping sub-regions characterized as too fast, too slow,too big, and too little are preferred in some embodiments.

Additionally, the phase of the timing signals is tracked to allowcorrections of T₀.

Allocation of Channel Control

The present invention further includes a unique system for allocatingcontrol of an established communication link between the originate andanswer modems (hereinafter designated A and B, respectively). Eachwaveform comprising the encoded ensemble of frequencies forms a packetof information.

Control of the transmission link is first allocated to modem A. Modem Athen determines the volume of data in its input buffer and transmitsbetween I (a minimum) and N (a previously determined maximum) packets ofdata as appropriate. The predetermined number N serves as a limit andthe end number of transmitted packets may be significantly less thanrequired to empty the input buffer. On the other hand, if modem A haslittle or no data in its input buffer it will still transmit I packetsof information to maintain communication with modem B. For example, theI packets may comprise the originate or answer comb of frequenciesdefined above with respect to FIG. 4 and the synchronization process.

Control of the communication link is then allocated to modem B whichrepeats the actions of modem A. Of course, if modem B transmits theminimum number, I, of packets it is confirming to modem A the vitalityof modem B.

There is no need for the limits N on the two modems to be the same, orto restrict them from being adaptable under modem control to obtainrapid character echo or other user oriented goals.

Hardware Implementation

FIG. 13 is a block diagram of a hardware embodiment of the invention.Referring now to FIG. 13, an electronic digital processor 120, an analogI/O interface 44, and a digital I/O interface 122 are coupled to acommon data bus 124. The analog I/O interface 44 interfaces the publicswitched telephone line 48 with the common data bus 124 and the digitalinterface 122 interfaces digital terminal equipment 126 with the commondata bus 124.

The following components are utilized in a preferred embodiment of theinvention. The analog I/O interface 44 is a high performance 12 bitcoder-decoder (codec) and telephone line interface. The interface hasaccess to RAM 132 and is controlled by supervisory microprocessor 128.The codec is a single chip combination of an analog to digitalconverter, a digital to analog converter, and several band pass filters.

The digital I/O interface 122 is a standard RS-232 serial interface to astandard twenty-five pin RS-232 type connector or a parallel interfaceto a personal computer bus.

The electronic digital processor 120, includes a supervisory processor128, a general purpose mathematical processor 130, a 32K by 16 bitshared RAM subsystem 132, and a read only memory (ROM) unit 133, coupledto an address bus 135.

The supervisory microprocessor 128 is a 68000 data processor subsystemincluding a 10 MHz 68000 processor and the 68000 program memory. The 32Kby 16 bit program memory consists of several low power, high density,ROM chips included in the ROM unit 133.

The mathematical processor 130 is a 320 digital signal microprocessorsystem (DSP) including a 20 MHz 320 processor, the 320 program memory,and an interface to the shared RAM system. Two high speed ROM chips,included in ROM unit 133, comprise the 8192×16 bit program memory.

The 320 system program memory includes programs for performing themodulation table look-up, FFT, demodulation, and other operationsdescribed above. The 68000 processor handles digital data streams at theinput and output, performs tasking to and supervision of the 320 signalprocessor and associated analog I/O, and performs self and system testas appropriate.

The invention has been explained with respect to specific embodiments.Other embodiments will now be apparent to those of ordinary skill in theart.

In particular, the ensemble of carrier frequencies need not be limitedas above-described. The number of carriers may be any power of 2, e.g.1024, or some arbitary number. Additionally, the frequencies need not beevenly spaced over the entire VF band. Further, the QAM scheme is notcritical to practicing the invention. For example, AM could be utilizedalthough the data rate, R_(B), would be reduced.

Still further, the modulation template need not be comprised of squares.Arbitrarily shaped regions surrounding the constellation points may bedefined. The tracking system was described where the squares in themodulation template were divided into four quadrants. However, a givenparameter may be tracked by tracking the difference in the number ofcounts in arbitrary regions defined about a constellation point.

Still further, a hardware embodiment including a supervisorymicroprocessor and a general purpose mathematical processor has beendescribed. However, different combinations of IC chips may be utilized.For example, a dedicated FFT chip could be utilized to performmodulation and demodulation operations.

Still further, the information units utilized in the above descriptionwere bits. However, the invention is not limited to binary system.

Accordingly, it is therefore intended that the invention can be limitedexcept as indicated by the appended claims.

What is claimed is:
 1. In a high speed modem communication system havingA and B modems coupled by a telephone line, a method for initiating asampling interval having a given time sample offset equal to thereciprocal of the sampling frequency, said method comprising the stepsof:generating an analog waveform at modem A including first and secondfrequency components at f₁ and f₂ ; transmitting said waveform frommodem A to modem B at time T_(A) ; adjusting the phases of said firstand and second frequency components so that their relative phasedifference at time T_(A) is equal to about 0°; detecting energy atfrequency f₁ at modem B to determine the estimated time, T_(EST), thatsaid waveform arrives at modem B; determing the relative phasedifference at modem B between said first and second frequency componentsat time T_(EST) ; calculating the number of sampling time offsets,N_(I), required for the relative phase of said first and second carriersto change from 0 to said relative phase difference; and changing themagnitude of T_(EST) by N_(I) sampling intervals to obtain a precisetiming reference, T₀.
 2. In a high speed modem communication systemhaving A and B modems coupled by a telephone line, a system forinitiating a sampling interval having a given time sample offset equalto the reciprocal of the sampling frequency, said systemcomprising:means for generating an analog waveform at modem A includingfirst and second frequency components at f₁ and f₂ ; means fortransmitting said waveform from modem A to modem B at time T_(A) ; meansfor adjusting the phases of said first and and second frequencycomponents so that their relative phase difference at time T_(A) isequal to about 0°; means for detecting energy at frequency f₁ at modem Bto determine the estimated time, T_(EST), that said waveform arrives atmodem B; means for determining the relative phase difference at modem Bbetween said first and second frequency components at time T_(EST) ;means for calculating the number of sampling time offsets, N_(I),required for the relative phase of said first and second carriers tochange from 0 to said relative phase difference; and means for changingthe magnitude of T_(EST) by N_(I) sampling intervals to obtain a precisetiming reference, T₀.